Subject: [NEXTEL-1] Article: EDGE modulation, how linearization improves amplifier performance

From: Marcel <[email-masked]>
Date: Sat, 04 Jan 2003 16:59:33 -0500

-------- Original Message --------
From: "Syed Maqbool Hassan <[email-masked]>" <[email-masked]>
Subject: Article: EDGE modulation, how linearization improves 

EDGE modulation, how linearization improves amplifier performance

By T. J. Fergus RF Design, Oct 1, 2002 
 
Historically, the global system for mobile Communications (formerly 
Groupe Speciale Mobile — GSM) was intended to make power amplifier 
design simpler. Typical GSM base transceiver system (BTS) power 
amplifier (PA) efficiencies are in the order of 40 percent. The 
simplified amplifier design requirement results from the use of 
Gaussian minimum-shift keying (GMSK). This modulation scheme 
maintains a constant RF envelope with only the phase of the signal 
varying with time.

With the introduction of high order modulation schemes for 2.5 and 
3G, the design challenges presented to the PA have increased 
significantly, for both base stations and mobile applications. The 
2.5G (enhanced data rates for GSM evolution — EDGE) and 3G (wideband 
code-division multiple access — WCDMA) standards have significant RF 
envelope variation due to the high order modulation schemes 
employed. They require linear amplification to minimize error-vector 
magnitude (EVM) distortion and limit adjacent channel power 
(spectral re-growth).

Clearly this requires highly linear power amplifiers operating in 
class A or AB, depending on the application. Typically, the required 
linearity is achieved by backing the PA away from compression. This, 
however results in poor efficiency (the base-station delivers less 
RF power and dissipates significant energy as heat). Typical 
efficiency for an EDGE PA is in the order of 15 percent (for 2.5 
percent EVM) for laterally diffused metal oxide silicon (LDMOS) 
transistors.

This article describes the design and implementation of a polar loop 
(closed loop phase and amplitude feedback system) to improve the 
efficiency of a gallium-arsenide field effect transistor (GaAs FET) 
power amplifier operating at 1.91 GHz.

This approach offers the following potential benefits when applied 
to an RF amplifier.

Significant improvement in efficiency and RF power handling for low 
distortions (For 2.5 percent EVM, improved power-added efficiency 
(PAE) from 13.9 percent to 18.1 percent and output power capability 
increased by 1.1 dB.

Minimum complexity due to novel architecture (direct comparison at 
RF without mix down to IF).

Gain and phase compensation for amplifier aging, tolerance and 
temperature.

The techniques presented here are aimed at base-station design where 
it is more beneficial to apply feedback in this manner. There is no 
reason why these techniques could not be used on mobile products 
given sufficient integration. Although applicable to most medium-
bandwidth modulation schemes the experiments presented are for GSM 
EDGE. A GaAs FET PA was chosen over LDMOS to show clearly the 
benefits of the polar loop in this application.

Feedback and distortion 
All amplifiers distort even at low levels. These distortions 
frequently become problematic at high drive levels. As an amplifier 
approaches compression, it imparts both phase and amplitude errors 
onto the signal passing through it. The result is AM to AM and AM to 
PM distortion, which generally gets worse as we approach 
compression. It is highly dependent on the amplitude of the signal 
and is directly related to the amplitude component of the modulation.

The effectiveness of any solution is dependent on the amplifier's 
characteristics, which should be determined in the first instance. 
If the amplifier exhibits a gentle compression characteristic then 
there is a good chance of improving efficiency by linearization.

The technique of using negative feedback to linearize power 
amplifiers is well known. Although the standard principles still 
apply, it is not a simple matter to implement negative feedback at 
RF frequencies when time delays in circuit elements are significant. 
The dangers of possible instabilities can be minimized using a 
thorough design process. This limits the bandwidth of such 
techniques — in effect they correct for the RF envelope distortion 
rather than the RF cycle itself. In other words it is the 
intermodulation products being corrected rather than the harmonics. 
This implies the need to demodulate the RF signal to determine the 
distortion artefacts. The approach adopted here does not need to 
demodulate the signal and does a direct comparison at RF.

For the purpose of this article, the application of feedback will be 
called "linearization" or "correction."

Linearization techniques 
The level of distortion in a typical GSM/EDGE PA is comparatively 
low — the devices used offer good linearity (normally LDMOS) at the 
onset. The use of open loop correction is unlikely to deliver the 
performance required to meet the requirements of EDGE in terms of 
EVM and adjacent channel power (ACP). Adaptive predistortion 
(baseband) is complicated and requires complete system integration. 
It cannot fully compensate for dynamic amplifier distortions and is 
approximate.

Closed loop systems can deliver significant performance advantage 
provided reasonable care is taken over their implementation. There 
are two main techniques used: feedforward and feedback. Feedforward 
requires the use of a second amplifier, which is an unwanted 
complication and requires careful calibration.

The polar loop is an elegant feedback system that applies the 
correction signals before the amplifier in the forward path using 
phase and gain modulators. This article discusses such a loop 
operating at 1.91 GHz. It further presents the mechanisms by which 
the loop operates with corresponding design compromises. Theoretical 
analysis is presented to the reader while the practical 
considerations of loop bandwidth, correction efficiency, stability 
and adjacent channel power are discussed in detail.

The polar loop 
The polar loop addresses both phase and amplitude distortion with 
two separate feedback loops. This implementation of the polar loop 
is novel in the sense that the phase and amplitude comparison are 
done at RF. Generally, polar loops require a mix down to an 
intermediate frequency or baseband itself (this eases the detection 
process). Recent advances in analogue ICs have made phase and 
amplitude detection simpler at RF frequencies.

The benefits of the device are used in the application of this novel 
polar loop. A simplified block diagram of the polar system is shown 
in Figure 1.

The theory of operation is simple. The phase and amplitude errors 
are detected by comparison between the reference and output signals. 
The amplitude error is derived from the difference between the 
reference and feedback signal envelopes. The detected phase error is 
the actual phase between RF reference and feedback cycles.

These errors are amplified (or integrated) and used to drive 
opposing modulators in the forward path. Therefore the loops attempt 
to minimize the amplifier distortions by cancelling them in the 
forward path.

A major problem is time delays in the system. In effect, we are 
comparing a reference (no delay) with a feedback signal (delayed by 
the forward path and amplifier and feedback circuits). It is 
possible to equalize this delay using a delay line in the reference 
path so that the comparison takes place at the right time.

The resultant error is now being applied to the forward path after 
the event has occurred! This fundamentally limits the correction 
bandwidth. The delay line can be omitted for narrow band frequency 
operation when the phase variation is smaller and can be 
accommodated by the loop. Generally, the modulation bandwidth is 
much lower than the RF carrier frequency so that the envelope and 
phase variations over the time delay in the circuits are minimal.

As a general rule of thumb, the correction bandwidth needs to be 
five times the modulating signal bandwidth. For example, assume we 
have a GSM signal with a bandwidth of 200 KHz. The correction loop 
bandwidth would need to be in excess of 1 MHz to offer a significant 
benefit.

Amplitude loop mathematics 
The following section uses mathematical analysis to predict the 
behavior of the polar loop.

Where:

Da(t) are the distortions generated in the amplifier:

Df(t) are the distortions due to the feedback path:

A is PA gain:

Ka is the amplitude modulator:

Af is the feedback amplifier:

Kd is the amplitude detector gain:

U(t) is the input signal:

Y(t) is the output Signal.

Next, performing the analysis:

Where F(t) == amplitude error signal.

Let K == Kd•Af•Ka, and therefore:

Hence, assuming ABK >> 1 and K >> 1, then:

Generally the phase loop can be analyzed in the same manner.

Conclusions drawn from analysis 
"B" sets the closed loop gain. Any distortion caused by the PA is 
reduced by the loop gain ABK. A loop gain of 20 dB should give a 
reduction of 20 dB in any intermodulation products falling within 
the loop bandwidth.

The feedback loop cannot distinguish between PA distortions and non-
linearities in the reference and feedback paths. It is important to 
ensure that the correction circuits add minimal distortion — 
applying correction to any system results in inherent distortion due 
to the correction loop itself. It can only improve matters if the 
distortion in the PA is much greater than that inherent in the 
closed feedback loop itself. This is a low pass effect and can be 
ignored outside the loop bandwidth.

Any distortion introduced in the forward path (due to modulators) 
will be significantly reduced by the feedback. It is, however, 
sensible to ensure that the distortion caused by the modulators is 
low otherwise we are correcting for the modulators rather then the 
PA.

The phase margin of the loop sets the maximum dynamic phase 
variations that can be corrected and still ensure stability. This is 
affected by variations in PA gain (assumed reasonably linear for 
EDGE) and the accuracy of the static phase shifter.

As the amplifier starts to distort, the additional phase distortion 
directly detracts from the phase margin of the loop.

Ideally, the phase and amplitude loops should be orthogonal. Any 
cross-coupling may detract from phase margin.

Practical considerations 
The results presented here are for a 3-stage lineup operating at 
1.91 GHz. The particular lineup for this discussion consists of the 
following devices: A NGA386 (Stanford Microdevices); a CMM1321 
(Celeritek) and a MGF0909 GaAs FET (Mitubishi).

This is a realistic scenario where the forward path modulators can 
be applied before the NGA386. The low signal levels mean that the 
forward path components add little distortion to the complete lineup 
(good linearity) and little power is wasted (insertion loss).

The DC power consumption of the PA is 12.5 W (10 VDC at 1.25 A 
controlled by active bias). The main polar loop components are an 
integrated amplitude modulator based on a 3 dB, 90° hybrid and an 
AD8302 gain/phase detector.

Each circuit was constructed then tested for performance. Critical 
issues for these devices were obtaining sufficient modulation 
bandwidth while maintaining minimal cross-coupling with the other 
loop. Typical modulation bandwidths achieved were in the order of 15 
MHz.

Initially the loops were designed using traditional theory to ensure 
stability with adequate phase margin. This set the maximum loop 
bandwidths that could be safely set without danger of oscillation 
(these were approximately 2 MHz wide). This represents the maximum 
performance for error correction of phase and amplitude and results 
in the best EVM improvements.

A switch was provided for each loop to select very low loop 
bandwidth — effectively turning the correction off but centering the 
loops. The amplitude loop was locked, then the phase using length of 
coax to center the loop at the midpoint.

Linearity 
Once locked and centered, the linearity of the closed loop was 
measured and is shown in Figure 3.

It is easy to see the effect of the amplitude correction loop on the 
lineup characteristic (the curves have been offset for clarity). The 
lineup manages a compression point of 37.6 dBm, which increases this 
to approximately 38.5 dBm when the linearizer is turned on. 
Effectively we have traded a soft knee characteristic for a more 
linear slope with a hard limit. This is a static test and does not 
verify the loop bandwidths necessary for modulated signals. However, 
the potential improvement is obvious.

Error vector measurement 
To verify the operation of amplitude and phase loops a modulated 
signal (EDGE) was used. An important parameter of the EDGE signal is 
the EVM, which gives a measure of the phase and amplitude distortion 
of the EDGE signal. The measured results are shown in Figure 4. The 
three traces show EVM for closed loop (polar loop on) open loop 
(lineup with phase and amp modulators in forward path) and lineup on 
its own.

The open loop system shows the polar modulators add little 
distortion to the lineup in open loop mode. With the polar loop on, 
the loop holds the EVM very low until it finally increases at 
approximately 34.8 dBm. The 0.40 percent EVM is the residual present 
in the test kit.

The potential gains are very good if the required system EVM is very 
low (roughly 6.0 dB more output power for 0.5 percent EVM). The main 
EVM improvement comes from the amplitude loop, however, the phase 
loop needs to be operational to achieve sub 1 percent EVM. The 
benefits decrease if higher EVM values are specified. For a target 
EVM of 2.5 percent we can obtain 1.3 dB more output power from the 
PA with the linearizer active.

ACP due to modulation 
The widest loop bandwidths and highest loop gains give the maximum 
benefit for EVM. However, this results in increased ACPM and 
wideband noise. This is an intuitive conclusion as the loop adds 
noise and distortion that spreads out within its bandwidth and 
appear at the PA output. In reality, there is a balance to be struck 
with loop bandwidth and gain to achieve the best compromise and that 
depends on the modulation, symbol rate and system specification.

The loop bandwidths were optimized for EDGE operation that gave the 
best EVM performance while still meeting ACPM specifications.

This gives the best EVM while meeting ACPM. The loops cannot be 
narrowed further, otherwise the loops will be too slow and will not 
be able to find lock in the ramp up period when frequency hopping. 
Figure 5 shows the close in ACPM plot for the amplifier producing 
+33.9 dBm.

It can be seen that the loop raises the modulation skirts however 
they still meet the requirement in the GSM specification 05.05. ACPM 
is affected almost entirely by the amplitude loop. In this 
situation, as the system starts to clip, the growth of ACPM is much 
greater for the linearized case. This can be easily understood by 
looking at the amplitude error signal. As EDGE has significant 
envelope variation, the amplitude loop tries to produce a faithful 
copy at the output.

As the input drive is increased there comes a point when the 
amplifier can no longer deliver the peak powers (hard clipping) and 
the amplitude loop clips.

Figure 6 shows a plot of the amplitude error signal — signal 
clipping on the envelope peaks can be clearly seen. (Note that a 
negative voltage gives max gain in this instance).

Clipping is a highly nonlinear action producing spectral spreading 
and poor ACPM. Clipping should not be allowed to occur in normal 
conditions.

The phase loop does not exhibit this behavior provided the range is 
sufficient and correctly centered. After reducing the loop gains and 
bandwidths, the EVM improvement is not as marked as before, reducing 
an uncorrected 7.5 percent down to 2.5 percent (the amplifier can 
now be driven harder and still meet ACPM specs). This represents a 
1.1 dB improvement in power over the basic lineup producing 2.5 
percent EVM. This is the trade off between EVM and ACPM.

Both loops eventually lose lock at low power levels. During these 
periods its important that the loops do not saturate, otherwise 
there will be an extended delay before the RF returns with the next 
TX burst. A simple diode arrangement can prevent saturation of the 
error amplifiers.

This system offers 55 dB of closed loop dynamic range (before phase 
loop fails).

Wideband noise 
The loop wideband noise performance is shown in Figure 7 for the 
amplifier (with linearizer on) delivering +33.9 dBm.

It can be seen that the noise floor increases for the closed loop 
case within the loop bandwidths. The effect is quite symmetrical, 
which is a good sign. Often, if the loop stability is marginal or 
there is loop cross-coupling between phase and amplitude loops, the 
wideband noise can have asymmetrical humps appearing above the 
floor. The performance here is good enough for a GSM BTS and does 
not display any signs of instability.

Hopping/static operation 
The speed of operation is a concern. For example, in GSM, the guard 
period between slots is 10 ìS. Therefore the loop needs to be able 
to attain lock in this period when frequency hopping between slots. 
Therefore this impacts on the bandwidth that the correction loop 
must achieve. The feedback loops must not be allowed to saturate 
when the RF signal is not present. Figure 8 shows the loop response 
to framed EDGE data (one slot on, 7 slots off).

In bursted mode, the loop lock time is good (less than 10 ìs), with 
EVM being maintained at 2.5 percent at the rated power. The falling 
edge is slow — the RF has ramped down and the amplitude drifts off 
due to offsets in the feedback and reference comparison signal.

Harmonics and filtering 
As proven by the earlier analysis, any distorting element in the 
feedback path will result in output distortion. There are some 
instances when there are needs for filtering. For example, harmonics 
of the amplifier could enter the feedback path and confuse the 
detection circuits without appropriate filtering. Any AM/PM 
distortion in this filter will distort the PA output.

In this instance a filter is not required due to the high linearity 
demanded by the application.

Conclusions 
This article discussed the design and implementation of a polar loop 
design using direct comparison at RF. The GaAs FET PA tested in this 
paper proved a suitable candidate for linearization with a good 
improvement in efficiency for a given EVM. However, similar results 
can be expected with any manufacturer's devices with similar 
parameters.

The measurements conducted showed the trade offs required to satisfy 
GSM EDGE 05.05 specification. Clearly this technique works well for 
narrow and medium modulation bandwidths (such as EDGE) and can 
achieve very low distortion performance acceptable for BTS 
applications. The implementation of the system minimizes the 
overheads associated with such linearization techniques while 
optimizing performance.

The application of closed loop polar feedback has been shown to 
increase the PAE of the GaAs FET amplifier. The potential benefit is 
more marked for lower EVM limits.

For 0.6 percent EVM, PAE increased from 5.2 percent to 14.7 percent. 
For 2.5 percent EVM, PAE increased from 13.9 percent to 18.1 percent 
with an increase in output power capability of 1.1 dB.

The performance improvement depends on many parameters including the 
amplifier linearity, amplifier delay, EVM, ACPM, modulation type, 
symbol rate etc. These need to be considered in addition to the cost 
overhead, additional power consumption, PCB area and general 
complexity. Only then is it possible to determine the business case 
for such a system. Undoubtedly, the obvious gains will be for high 
power BTS. With suitable integration and the ever-increasing demand 
for battery lifetime this technique may prove useful on mobile 
handsets in the future.

References 
GSM 05.05 "Radio Transmission and Reception"

S. C, Cripps "RF Power Amplifiers For Wireless Communications" 
Artech House 1999 ISBN 0-89006-989-1

P B Kenington "High Linearity RF Amplifier Design" Artech House 2000 
ISBN 1-58053-143-1

V Petrovic and W Gosling "Polar Loop Transmitters" Elec. Letters, 10 
May 1979, Vol 15, pp 286 - 287

Analog Devices AD8302 Datasheet.

About the author 
The author would like to thank all of his colleges in the wireless 
technology practice for their assistance and helpful comments during 
this research.

Tim Fergus is a consultant working at the Wireless Technology 
Practice of PA Consulting Group, Melbourn, Hertfordshire, United 
Kingdom. He holds a 1st Class B.Eng in Electronic and Electrical 
Engineering from Brunel University, Uxbridge, Middlesex, in 1993. 
Tim has worked in consultancy for more than six years specializing 
in RF and high speed analogue electronics. He has worked on a broad 
range of communication systems from bespoke formula 1 telemetry 
systems through to digital cellular systems (handset and base 
station) such as GSM, DECT and PCN. He also has worked for a wide 
range of clients providing both system and circuit level design 
expertise. He can be contacted at [email-masked]
 
Source:
http://rfdesign.com/ar/radio_edge_modulation_linearization/index.htm

Best regards
Syed Hassan

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