Subject: [NEXTEL-1] A DSSS-based wireless short range data-link

From: [email-masked]
Date: Wed, 30 Jul 2003 01:21:29 -0400

A DSSS-based wireless short range data-link

Source: International Journal of Electronics and Communications
Publication date: 2003-01-01
Arrival time: 2003-07-29

Dedicated to Professor Peter Russer on the occasion of his 60th birthday
Abstract: The system concept for the physical layer of a wireless high speed
data-link is introduced for local-area-ad-hoc-networks in mobile applications. The
high speed data-link is based on a direct sequence spread spectrum system. The
dynamic losses in a incoherent receiver at 24.1 GHz are investigated.
Keywords: DSSS, Mobile, Wireless, Demodulation, Signal processing

1. Introduction

There is an increasing demand for low cost high speed wireless ad-hoc-networks in
mobile applications for multi-media applications in mobile systems as e.g. WPAN
[1] and WLAN-systems [2]-[4] being already available in cars, as well as inter
vehicle communication [5],[6]. For mobile ad-hoc-networks there are special
demands to be fulfilled like high dynamics, channel separation and high immunity
against external interferers. Code Division Multiple Access- wireless systems have
proven to fulfill these demands well, if there is enough bandwidth available
[7]-[11].  The direct sequence spread spectrum (DSSS) of the CDMA-systems with
high chiprate enables channel separation, a high immunity against external
interferers and the transmission with low average emission level.

In this paper we introduce the physical layer of a CDMA-DSSS wireless short range
data-link where a flexible DSSS system is used for the data transmission of
variable data rates. The main requirement for the data-link is to achieve a
reliable (p^sub e^ < 2 [middot] 10^sup -4^) wireless bi-directional data
transmission with a variable data rate of up to 12 Mbit/s.  The transmission
should not be locatable from far distance and therefore the operation range should
be limited to 10 m. Since the transmission is appropriate for sensitive data, it
should be secure against jamming, interference and interception. The considered
wireless link represents the physical layer of a wireless network with ad-hoc
abilities. An ad- hoc wireless network is a collection of wireless mobile hosts
forming a temporary network without the aid of any established infrastructure or
centralized administration. This type of network is of great importance in
situations where it is difficult to provide the necessary infrastructure e.g. in
mobile wireless systems, but it is a challenging task to enable fast and reliable
communication (and link establishment) within such a network. A small current
consumption must be achieved in order to prevent a strong load of the batteries in
the mobile system.
In order to fulfill all these requirements, a special system concept has been
compiled. The immunity against jamming and eavesdropping are to be achieved by a
DSSS-transmission. In such a system data can be hidden in the background noise by
spreading its bandwidth with a pn (pseudo random noise)-code and transmitting the
resulting signal at a low average power. Because of its low power level, the
transmitted signal has a low probability of being intercepted by an unauthorized
participant.  Furthermore the data can only be demodulated by a receiver which
knows the applied pn-code. The transmitting power should be adjustable over an
automatic control loop so that during a reliable transmission the position of the
transmitters cannot be located (range < 10 m). By this "minimum power consumption"
also battery operation is possible.

2. System architecture of the wireless data-link

In the following the system concept is shown (Fig. 1). The homodyne data-link is
divided into a digital signal processing unit and an RF-frontend.  Since the
concept of the data-link must be flexible with respect to variable data rates,
also the baseband signal processing must be adaptable to variable data rates.
That's why for this stage of the investigation a field programmable gate array
(FPGA) is chosen for the realization of the digital part since FPGAs are freely
programmable and hence, a change of the data rate and the signal processing will
not need a change of the hardware. An essential question for the RF-part is the
operating frequency band. The frequency band determines also the antenna size and
the available bandwidth. For the data-link the 24.1 GHz ISM band has been selected
which is license-free available. The available bandwidth is 250 MHz and hence, the
use of small antennas with adaptable gain and the application of digital
beamforming is possible. While in the very common ISM frequency bands at 2.4 GHz
and 5 GHz there are lots of services leading to possible interference's, the
probability of inter-ference's are more improbable at 24.1 GHz from today's view.
In addition, the freespace path loss is also higher at 24.1 GHz. Therefore the
emitted signals do not harm other systems and the own system can not be located
easily from far distances.
Fig. 1. System architecture of the physical layer of the wireless short range
data-link.

The digital circuit consists of two parts - the transmitting path and the
receiving path. In the transmitting path, the binary data d(t) is multiplied with
the pn-code c(t), to produce the transmitted baseband signal g(f) = d(f) [middot]
c(t), In order to be able to differentiate between the terms bit and bit rate of
the pn code we designate it as chip and chip rate [8]. For a number of n chip in
one code sequence the chip rate of the pn-code c(t) is n-times greater than the
data rate of d(t). The effect of multiplication of d(t) with the pn-code is to
spread the baseband bandwidth of d(t) to the baseband bandwidth of the pn-code
c(t) witch is also much greater [11] and hence less sensible against
interferences. The digital part must first synchronize the incoming data with the
pn- code. This means that the starting time and stop time of the pn- code agree
with the starting time and the stop time of the data bits. After this
synchronization the multiplication is be performed. The chip rate C is a multiple
of the data rate D with C = n [middot] D where n is the number of chips in the
pn-code sequence. Hence the resulting transmitted baseband signal g(t) has got the
chip rate C of the pn-code. This signal, produced in the digital part, is then
modulated in the RF-part.

In order to realise a BPSK-based transmission an RF-frontend with stable RF-source
is required where the carrier frequency of the VCO is controlled via a phase
locked loop which yields synchronization with the clock frequency. For the clock a
stable quartz oscillator is used. After the demodulation of the received signal
into the baseband the signal is transferred into the FPGA via an A/D- converter.
In the receiving path of the digital circuit the received signal has to be
despreaded. In order to despread the received signal, it has to be correlated with
the same synchronized pn-code (the pn-code used in the transmitter). This
operation is called despreading, since the effect is to undo the spreading
operation at the transmitter [7]. Therefore a correlation receiver has been
implemented into the FPGA. The received signal is then correlated with the
pn-code. A decision logic derives then the transmitted data signal from the
correlation. In the receiving path the amplitude of the received signal is
determined by an A/D-converter after the demodulation. The output of the
A/D-converter is also routed to the FPGA. Using a special closed loop control
(witch is adapted to the characteristic of the output amplifier) the power of the
transmitter is controlled.
In what follows the gain of the correlation receiver due to the SS-system is
investigated in dependence to the frequency deviation between the carrier
frequency of the received signal and the frequency of the RF-source used for
demodulation in an incoherent receiver. The correlation gain is used to increase
the signal noise ratio of the spread spectrum system and is hence an essential
value of the concept. After the received signal r(t) is transferred to the digital
level in the FPGA the signal is sampled with a sampling rate of four time steps
per chip using four parallel correlators. In Fig. 2. the signal is shown for the
case of a pn-code of length n = 7.

Because of the incoherent transceiver there is a frequency shift of
[Delta][omega] between the carrier frequency of the received signal and
the frequency of the VCO used for down-conversion. This signal is
correlated with the pn-code sequence c(t). The sequence length is n, the
duration of a chip is [Delta]t = 1/C and the pn-code duration

Fig. 2. left side: received signal after demodulation in a coherent receiver;
right side: received signal after demodulation in an incoherent receiver for a
relative frequency deviation of 0.01 % at 24.1 GHz and a data rate of 12 Mbit/s.
The time shift of [tau] = 0 in the case of synchronization of the pn-code with the
pn-code in the received signal which is ensured via a tracking algorithm realized
in the FPGA. In Fig. 3. the correlation function is shown for the case of a
frequency deviation of 0%, 0.01 % at 24.1 GHz at a data rate of 12Mbit/s.

Now we want to consider the worst case for the correlation gain in
dependence of the frequency shifts [Delta][omega]. The gain can be
obtained from the maximum values of the correlation function which occur
at [tau] = 0. Following [8],[9] we obtain for the correlation gain
G([Delta][omega])= 10 1g | R^sub c^(0, [Delta][omega]) | (4)

Fig. 3. Correlation after demodulation in an incoherent receiver for arelative
frequency deviation of 0.01 % at 24.1 GHz and a data rate of 12 Mbit/s, comparison
with the auto-correlation function R^sub a^.
In the ideal case, for the auto-correlation of the pn-code, the gain is G^sub
max^ = 10 1g(n) where n is the number of chips in the code. In Fig. 4. the
absolute value of the correlation function is shown for the case of a
frequency deviation of 0%, 0.01%, 0.02% and 0.03% at 24.1 GHz at a
data rate of 12 Mbit/s. As can be seen, even in a coherent receiver, for the
frequency deviation of 0%, the maximum values of the correlation function
and hence the correlation gain G are 16% lower than the ideal
auto-correlation value. This is because of the bandpass filtering of the
signal with a bandwidth of 250 MHz. We see that for a noncoherent
receiver with a frequency deviation of 0.01 %, we still obtain a good
correlation gain which is only 7% lower than for the case of a coherent
receiver. In order to obtain a general behavior of the Gain dependent to the
fre-

Fig. 4. Absolute value of the correlation ...^sub cm^ with a relative frequency
deviation of m[middot]0.01% for m = 0, 1, 2, 3 at 24.1 GHz and a data rate of 12
Mbit/s; comparison with the auto- correlation function R^sub a^.

Fig. 5. Basic structure of the correlation receiver.
from the results of the individual receivers and assigns these to d^sub a^(t). The
complete correlation receiver is then implemented using the hardware description
language VHDL. The circuit is flexibly developed so that without large expenditure
other codes can be used. The threshold values are also adjustable flexibly. Thus
the circuit can be extended to an automatic threshold level control as it is
discussed in [14].

For the wireless data-link two advantageous spreading codes have been designed,
having the length of 8 chip and 12 chip respectively. These codes are able to
increase the signal noise ratio considerably by an order of magnitude but they
cannot be used for the channel separation of more than three participants if we
are using small codes with up to 12 chips.  The small number of chips in the codes
is restricted by the ratio between the allowed bandwidth and the data rate. In the
system concept the channel separation is supported by an additional TDMA protocol.

3. Realization and measurements

The entire circuit of the data-link has been described in VHDL, simulated and
implemented into a commercially available FPGA. In order to keep the space
requirement low a relatively small FPGA (50 000 SystemGates) is used. Fig. 6 shows
the physical space requirement of the circuit on the FPGA after the
implementation. The space requirement corresponds to 63% of the FPGA capacity.

Fig. 6. Physical space requirement of the circuit on the FPGA; covered space
marked by the dots.

Fig. 7. Demonstrator of the CDMA-DSSS-transceiver and possible positions for
integration into a car.

Fig. 8. Coded video data before spreading and after despreading.

In the actual demonstrator two transceiver modules are realized. In order to keep
the space requirement and the weight small, patch- antennas of size 4 [middot] 4
mm^sup 2^ are used. The modules are small and hence they can be easily integrated
into a car for demonstration of their function (Fig. 7). Since the digital circuit
is realised in an FPGA and completely described in VHDL, it can be implemented
also into other FPGA's [15].  The signal processing is performed completely in a
field programmable gate array which is base of a flexible design and also offers
the possibility of flexible re-programming steered by the protocol layer. The
capability of the wireless data-link has been tested in a video transmission.
MPEG-2 coded video signals has been transferred at a data rate of 2.5 Mbit/s. In
order to guarantee a reliable transmission, the error rate must be better than
p^sub e^ = 2 [middot] 10^sub -4^. These video data are transmitted and decoded
correctly. Fig. 8. shows a comparison of the received and reconstructed data at
the output of the data-link (see the below signal) and the coded data at the input
of the data-link.

As can be seen in Fig. 8, the entire time for the digital signal processing
(transmitting and receiving path together), RF- modulation/demodulation and the
transmission amounts to only 400 ns. That is very favourable for real time
applications like video transmissions. To demonstrate the maximum data rate a 12
Mbit/s signal has been also transferred. All measurements were accomplished
without jamming j(t). The concept shows that with relatively simple and cheap
components a very small wireless DSSS demonstrator for the transmission of up to
12 Mbit/s can be developed and investigated.

4. State of the art

In the following we discuss different wireless communication systems for short
range communication with respect to the demands described in the introduction. In
the last years numerous wireless short range communication systems have been
developed and standardized for different applications. Table 1 gives an overview
of the different short range communication systems including the investigated
data-link and its characteristics [4].

Bluetooth can manage a small number of low-cost point-to-point radio links between
Bluetooth-enabled devices. It has the required ability to establish
ad-hoc-networks and regulates the transmitting power by demand. It is operated in
the license-free 2.4 GHz ISM frequency band, and applies Frequency Hopping Spread
Spectrum (FHSS) with up to 1600 hops/sec for transmitting data at a rate of 1
Mbit/ s over the air. Since the transmission and the transmitted power is burst
oriented it can be located more easily. Although it has excellent network and
system capabilities it achieves low data rates compared to the above described
requirements.  From today's point of view a payload data rate of 12 Mbit/s can not
be expected in the next years development.

DECT (Digital Enhanced Cordless Telecommunications) is also a established system
and doing well as a wireless telephony system. It achieves a data rate of 1.152
Mbit/s. The main benefit of DECT in comparison to the other standardized systems
in Table 1 is the frequency band. It operates in its own dedicated frequency band
which ranges from 1.88-1.9 GHz (in Europe) where there are applied only few
services which could interfere the function as it is possible in the 2.4 GHz or 5
GHz ISM bands. A further difference is the transmission procedure. DECT makes use
of a multi carrier transmission and works without a spread spectrum transmission.
A data rate of 12 Mbit/s can not be expected in the next years development. This
advantageous dedicated frequency band is relatively narrow and the transmission
and the transmitted power is burst oriented. Therefore the system is easier
locatable.

The HomeRF system has good network characteristics and works in the 2.4 GHz
frequency band. The data rate is at present 1.6 Mbit/s and at the second
generation (HomeRF 2.0) even 10 Mbit/s. This system uses a FHSS with 50 hops/sec.
In the next years development the data rate is expected to be increased up to 20
Mbit/s. But this high-speed system is not available at present since it is under
development. Since the transmission and the transmitted power of HomeRF is burst
oriented it can be located more easily.

Table 1. Overview of the different wireless short range communication
systems

Since a low transmitting power is required a DSSS system is to be preferred. The
IEEE 802.11b is also an established standard with flexible network capabilities
and doing well as a WLAN system. It achieves nearly the required data rate and
uses DSSS for the transmission in the 2.4GHz band. Several data bits determine the
choice of the spreading code. That means that the pn-code transports also data
information. That leads to a lower correlation gain for the DSSS. Therefore the
system achieves the data rate of 11 Mbit/s only in relatively reliable radio
channels. If the channel quality decreases it reduces the data rate up to 1 Mbit/s
which is to low for the requirements of data-link although this system fulfills
most of the requirements mentioned in the introduction.

In the table one can see that some of the WLAN systems achieve the required data
rate of 12 Mbit/s easily (IEEE 802.11a and HiperLan/2) [3],[12]. These OFDM based
high-speed systems operate in the 5 GHz ISM band and reduce their data rates from
54 Mbit/s up to 6 Mbit/s if the channel quality decreases. In contrast to DSSS
systems these systems transfer on several channels simultaneously. Therefore these
systems are locatable more easily than DSSS systems.

Although every of these powerful established systems fulfills some of the above
described requirements e.g. ad-hoc-abilities or data rates, none satisfies all of
these special demands. Table 1 shows that in the very common ISM frequency bands
at 2.4 GHz and 5 GHz there are lots of services leading to possible
interference's. The system at 24.1 GHz generates less interference towards other
distant systems because the path loss is relatively high at this special frequency
and there are less services expected from today's point of view. The system must
be also flexible in changing the data rate and the code-length by demand and under
consideration of the higher bandwidth at 24.1 GHz to achieve the best possible
correlation gain. The higher carrier frequency also offers the use of digital
beamforming with smart antennas of acceptable size. The wireless data-link at 24.1
GHz yields a flexible platform for data transmission fulfilling all of the above
described requirements e.g. a flexible data rate, a flexible code-length, low
emission power and low locateability.

5. Conclusions

The system concept of a wireless DSSS data-link has been investigated enabling the
transmission of variable data rates up to 12 Mbit/s with high reliability. In
comparison to already available higher speed ad-hoc networks which use e.g. 2.4
GHz carrie\r frequency, this concept is based on a 24.1 GHz carrier frequency
which allows a higher bandwidth of 250 MHz for the SS-signal. The digital raw data
is spread by an advantageous 8-chip code. Optionally 12-chip-codes can be used for
higher correlation gain and for higher channel separation. The wireless DSSS
data-link has been realized and tested in a video transmission path after
performing a digital data compression. The data compression required a bit error
rate of less than 2 [middot] 10^sup -4^ which could be fulfilled easily by the
system.
Acknowledgement. The authors would like to acknowledge the support of H. Ni, S.
Mayer and J.F. Luy.
(C) Urban & Fischer Verlag
http://www.urbanfischer.de/journals/aeue
References
[1] Nusser, R.; Pelz, R. M.: Bluetooth Based Wireless Connectivity in an
Automotive Environment. IEEE VTS Fall VTC2000, Boston, 4 (2000), 1935-1942.
[2] Sikora, A.: Wireless LAN Protokolle und Anwendungen. Addison-Wesley, 2001.
[3] IEEE 802.11a Part 11: Wireless LAN Medium Access Control (MAC) and Physical
Eayer (PHY) specifications - High-speed Physical Layer in the 5 GHz band +
Extension in the 2.4 GHz Band. IEEE, 1999.
[4] Merkle, A.; Terzis, A.: Digitale Funkkommunikation mit Bluetooth.
Franzis Verlag, 2002.
[5] Lindenmeier, S.: High speed wireless data links for mobile applications.
European Conference on Wireless Technology, Milan Sept.  2002, 19-22.
[6] Lindenmeier, S.; Mayer, S.; Knopfle, D.; Luy, J.-F.: Communicating Near Range
Sensor System for Automotive Application. MTT-S Workshop Digest, Seattle,
2-7.6.2002.
[7] Lee, S. L.; Miller, L. E.: CDMA Systems Engineering Handbook.
Artech House, 1998.
[8] Goiser, A. M. J.: Handbuch der Spread-Spectrum Technik. Springer Verlag Wien,
1997.
[9] Viterbi, A. J.: CDMA Principles of Spread Spectrum Communication.
Addison Wesley, 1995.
[10] Polydoros, A.; Weber, C.: A unified Approach to Serial Search Spread-Spectrum
Code Acquisitation. IEEE Trans. on Communications, COM-32(5) (1984).
[11] Finger, A.: Pseudorandom Signalverarbeitung. B. G. Teubner Stuttgart, 1997.
[12] IEEE Std 802.11b-1999 Part 11: Wireless LAN Medium Access Control (MAC) and
Physical Layer (PHY) specifications: Higher-Speed Physical Layer Extension in the
2.4 GHz Band. IEEE, 1999.
[13] Proakis, J. G.: Digital Communication. 4th edition McGraw- Hill, 2001.
[14] Glisic, S. G.: Automatic Decision Threshold Level Control in Direct-Sequence
Spread-Spectrum Systems. IEEE Trans. on Communications 39(2) (1991).
[15] Asheden, P. J.: The Designer's Guide to VHDL. Morgan Kaufmann Publishers,
1996.
Received August 23, 2002. Revised November 12, 2002.
DaimlerChrysler Research Center, Wilhelm-Runge-Strasse 11, 89081 Ulm, Germany.
Phone: +49/731505-2024, Fax: -4102.
Correspondence to S. Lindenmeier.
E-mail: [email-masked]

Stefan Lindennieier was born in Munich, Germany, in 1967. He received the Dipl.
Ing. degree in 1994, the Dr.-Ing. degree in 1996 and the Dr.-Ing.  habil degree in
2000 in electrical engineering at the Technical University of Munich, Germany.
From 1994 to 1996 he has been with the Ferdinand Braun Institut in Berlin,
Germany, where he was working on numerical techniques for the analysis of planar
microwave circuits. In 1996 he joined the department of high frequency techniques
at the Technical University of Munich, Germany as an assistant professor where he
worked on antenna design, EMC, and radar techniques. In 1997 he was recipient of
the Dr.  Georg Spinner high frequency award. Since July 2000 he is with the
DaimlerChrysler Research and Technology Center in Ulm, Germany, where he is
director of a research group for microwave communication and near range sensors.
His current research interests are the development of new mobile communication
systems, signal processing, antenna design and near range radar sensors for
automotive applications. Stefan Lindenmeier is member of the ITG FA. 9.1. He is
author of more than 60 scientific publications in the field of high frequency and
communication techniques.

Anestis Terzis was born in Heidenheim, Germany, on June 17, 1978. He received the
Dipl.-Ing. (FH) degree in electrical engineering on February 2002 from the
University of Applied Sciences Ulm, Germany. Afterwards he wrote a book concerning
the communication system Bluetooth. In October 2002 he joined the department for
Microwave Technology at the DaimlerChryslcr Research Center in Ulm, Germany. His
research interests in-elude wireless short range communication systems, simulation
and digital signal processing with VHDL. Anestis Terzis is member of VDE and VDI.

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